The present invention relates to the field of spread spectrum communications and more particularly to direct sequence spread spectrum transceivers.
Spread spectrum refers to a type of signal modulation in which the bandwidth required for transmission greatly exceeds the information bandwidth. Motivation for spread spectrum signals is based on the following: (1) the ability to reject intentional and unintentional jamming by interfering signals so that information can be communicated; (2) low probability of interception or detection since the power in the transmitted wave is xe2x80x9cspreadxe2x80x9d over a large bandwidth or frequency extent; (3) message privacy since the signals cannot be readily demodulated without knowledge of the code; (4) tolerance of multipath reception; and (5) the ability to send many independent signals over the same frequency band.
One generic type of spread spectrum signal and the most often used is a direct sequence spread spectrum (DSSS) signal. In direct sequence modulation, each bit of an information-bearing signal, the information desired to be transmitted which is inherently random or at least unknown to the receiver and contains the information to be communicated bit by bit, is modulated by a higher frequency, pseudorandom code signal. Direct-sequence spread spectrum is commonly achieved by directly modulating a carrier with a high rate digital pseudorandom spreading code. For example, digital data at a much lower rate is modulo-2 added to a pseudorandom code. Modulo-2 addition uses symbols consisting of 0 or 1 and is equivalent to multiplication with symbols consisting of xc2x11""s. Both the data and the pseudorandom code are multiplied by a carrier. This modulation method is often referred to as phase shift keying because the multiplication of data, code and carrier result in shifts of the carrier by 180 degrees. A bit value of one produces no phase shift and a bit value of minus one produces a 180 degree phase shift.
A generic DSSS signal can be represented as
si(t)=cos(xcfx890t+di(t)+m(t))xe2x80x83xe2x80x83(Eq. 1)
where xcfx890 is the carrier frequency, di(t) is the phase modulation due to a set of i data symbols, and m(t) is the phase modulation due to the spreading sequence. For conventional binary phase shift keying DSSS signaling this may be simplified to
si(t)=di(t)m(t)cos(xcfx890t).xe2x80x83xe2x80x83(Eq. 2)
Now di(t) represents the data waveform consisting of the binary symbols xc2x11, and m(t) represents the pseudorandom code also consisting of the binary symbols xc2x11. For binary phase shift keying signaling, the combination of code modulo-2 added with data will produce 1 of 2 symbols:
s1(t)=cos(xcfx890t)xe2x80x83xe2x80x83(Eq. 3)
s2(t)=cos(xcfx890+xcfx80)xe2x80x83xe2x80x83(Eq. 4)
These waveforms are called xe2x80x98antipodalxe2x80x99 signals since s1(t)=xe2x88x92s2(t). This is shown vectorially in FIG. 1. FIG. 1 shows an I or inphase-axis at 102 and a Q or quadature phase-axis at 101. The signals s1(t) and s2(t), respectively are shown along the positive inphase-axis at 103 and the negative inphase-axis at 100. The signals are antipodal by virtue of the fact that they are of equal magnitude and are 180 degrees apart in phase.
If the signal si(t) is squared, which is done when a simplified receiver is used, as is described below, and the trigonometric identity cos2xcex1=xc2xd+xc2xd cos 2xcex1 is applied, the following results:
[si(t)]2=di2(t)m2(t)cos2(xcfx890t)xe2x80x83xe2x80x83(Eq. 5)
then, for i=1 or 2 for binary signaling as in (3) and (4) above,
xe2x80x83s12(t)=cos2(xcfx890t+xcfx80)=xc2xd+xc2xd cos(2xcfx890t),xe2x80x83xe2x80x83(Eq. 6)
and
s22(t)=cos2(xcfx890t)=xc2xd+xc2xd cos(2xcfx890t).xe2x80x83xe2x80x83(Eq. 7)
The result is a direct current component and a sinusoid at twice the carrier frequency, reduced by 3 dB. A power reduction of xc2xd results in a 3 dB loss. That is, 10 log (xc2xd)=xe2x88x923.01 dB. The resulting waveform is devoid of both pseudorandom code and data components. This is a commonly known result of squaring binary phase shift keying antipodal signals. Traditional DSSS receivers, therefore, have not employed a mathematical squaring operation at the receiver because the antipodally modulated data would be eliminated in such operation.
FIGS. 2 and 3 show generally prior art arrangements of a DSSS transmitter and receiver, respectively. The transmitter of FIG. 2 shows a binary bit data sequence 200, a pseudorandom spreading code 201 both combined by a mixer 202. The signal is then modulated using a binary phase shift keying modulator 203 onto a carrier wave 204. The signal is then transmitted at 205. The prior art receiver of FIG. 3 receives the signal at 301. Using a stored reference, the pseudorandom code at 303 is eliminated by the mixer 302 and the transmitted data signal is recovered and then demodulated using a binary phase shift keying demodulator 304. The binary bit data signal is then outputted at 305 to be used in the specific application.
In traditional DSSS systems, for the receiver to function, a replica of the pseudorandom code must be generated in the receiver and must be brought into time and frequency alignment with the transmitted pseudorandom code. High implementation costs are associated with such receivers. A squaring loop is then used to recover the carrier while a parallel path retains the data for coherent demodulation. Their function is to track the carrier and thereby allow phase-coherent demodulation of the data, that is, multiplication by a signal exactly in phase with the received carrier.
One of the disadvantages of using DSSS is the cost of implementation due primarily to the difficulty in achieving synchronization with the high rate pseudorandom spreading sequence. Despreading of a DSSS signal without knowledge of the pseudonoise (pseudorandom) spreading sequence or code synchronization has been previously described by a technique known as xe2x80x9cblind despreadingxe2x80x9d. This is achieved with knowledge of key signal parameters by either the xe2x80x9cautocorrelationxe2x80x9d or xe2x80x9ccyclic autocorrelationxe2x80x9d methods.
The present invention provides a method and device for blind despreading that requires only knowledge of carrier frequency and signal rate and is made possible by the use of novel non-antipodal phase shift keying phase modulation. This provides the advantages of both low energy density and low implementation costs. Low energy density is useful for applications such as space based systems that are required to maintain low power spectral densities on earth to reduce co-channel interference and to satisfy regulatory constraints. Implementation costs may be significantly lowered since this technique requires no embedded reference code or synchronization at the receiver. This suggests a variety of wireless applications where the advantages of spread spectrum modulation are desired and where numerous low cost receivers are required such as for computer local area networks (LAN), emergency alerting devices, and space-based broadcast systems of all types.
The present invention comprises a reception simplified, reduced cost direct sequence spread spectrum transceiver device and method. The transmitting portion of the transceiver antipodally modulates the pseudorandom code and non-antipodally modulates the data onto a carrier wave. The receiving portion of the transceiver performs a mathematical squaring function on the received signal combination of pseudorandom code and data. Squaring the pseudorandom code removes the code with the data remaining in the absence of the pseudorandom code. The device and method of the invention eliminates the need for storing or generating a local reference for the spreading code or the need for any knowledge of specific signal parameters at the receiver, greatly simplifying receiver circuitry. Additionally, the device and method of the invention eliminates the synchronization constraint between the transmitted data and spreading code of conventional systems allowing receiver operation at low energy density in receiver apparatus and low operational cost.
It is therefore an object of the present invention to provide a reduced cost spread spectrum transceiver device and method that provides simplified reception.
It is another object of the invention to provide a spread spectrum transceiver device operable in the absence of a stored reference at the receiver.
It is another object of the invention to provide a spread spectrum transceiver device operable in the absence of specific signal parameter knowledge at the receiver.
It is another object of the invention to provide a spread spectrum transceiver device and method employing non-antipodal phase shift keying modulation.
It is another object of the invention to provide a spread spectrum transceiver device and method for that allows for M-ary phase modulation data schemes.
It is another object of the invention to provide a spread spectrum transceiver device and method that eliminates the synchronization constraint between data and spreading code of conventional spread spectrum systems.
It is another object of the invention to provide a spread spectrum transceiver device and method that has a lower energy density in receiver apparatus than conventional systems and can be implemented at lower cost.
These and other objects of the invention are described in the description, claims and accompanying drawings and are achieved by a despreading simplified, receiver circuitry-minimized direct sequence spread spectrum transceiver device comprising:
a binary bit data interfacing circuit;
a binary bit pseuodrandom code generating circuit;
a non-antipodal modulation preprogrammable digital synthesis phase modulating device communicating with said binary bit data interfacing circuit and outputting a binary bit data sequence of preselected frequency;
a binary bit pseudorandom code and binary bit data sequence mixing device, said output of said mixing device modulated onto a spread spectrum carrier wave;
an operator determinative frequency translating circuit communicating with said mixing device;
a mathematical squaring function spread spectrum receiving circuit, said receiving circuit receiving output from said operator determinative frequency translating circuit and performing a mathematical squaring function thereon, said squaring function eliminating said binary bit psuedorandom code and preserving said binary bit data sequence; and
said mathematical squaring function capability receiving circuit including an operator determinative binary bit data demodulating circuit.